High-frequency traveling wave field-effect transistor

ABSTRACT

A method of improving the performance of a traveling wave field-effect transistor operated at frequencies in the microwave range or above the microwave range comprising the steps of forming a depletion region beneath a gate electrode wherein, in a plane transverse to the direction of signal propagation, a depletion region edge has a first end portion located between the gate electrode and a drain electrode and a second end portion located between the gate electrode and a source electrode; and separating the depletion region edge from the drain electrode. Further improvements in the operation of the TWFET include adjusting the first end portion of the depletion region edge to be closer to the gate electrode relative to the distance between the second end portion of the depletion region edge and the gate electrode, controlling an effective conductivity of a semiconductor of the traveling-wave field effect transistor, and setting the length of the gate electode at about one micron.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is continuation-in-part of allowed U.S. patentapplication, Ser. No. 08/275,999, U.S. Pat. No. 5,627,389, filed Jul.15, 1994 of common title and inventorship as the present application.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to field-effect transistors, and in particular totraveling wave field-effect transistors (TWFETs) for use at frequenciesin the microwave range and above the microwave range.

2. Description of the Prior Art

The prior art concepts for the design of high frequency semiconductortransistors, in particular field effect transistors (FETs), haveobtained limited improvements in transistor performance in the aspectsof signal power output and signal power gain at these high frequencies.Previous attempts to improve the performance of FETs for operation atthese high frequencies have included the use of TWFETs. U.S. Pat. No.4,065,782 presents the TWFET as an FET in which some of the FETselectrodes are adapted for use as coupled transmission lines. Thisconcept was extended in U.S. Pat. No. 4,675,712, which presents a TWFETwith a drain electrode that includes a meander segment. In U.S. Pat. No.4,733,195, a TWFET is designed with unmatched termination impedances onthe input and output transmission lines.

A TWFET comprises parallel elongated source, gate and drain electrodesdisposed on a substrate of semiconductor material with an active channelconnecting the electrodes. The direction of signal propagation is alongthe axis of these parallel electrodes. In the plane transverse to thisdirection of signal propagation, i.e. the cross-section of the TWFET,the arrangement of the electrodes and channel form an FET. In otherattempts to improve the performance of TWFETs at high frequencies, thiscross-section of the TWFET has been modified. For example, in U.S. Pat.No. 4,297,718 the TWFET's cross-section is designed as a vertical FET.In U.S. Pat. No. 4,587,541, the TWFET's cross-section is designed asdual-FETs.

All of these devices, however, achieve moderate increases in signalpower output and signal power gain, particularly at high frequencies.

SUMMARY OF THE INVENTION

It is therefore a primary object of the present invention to providesignal power gain in a TWFET, operated at frequencies in the microwaverange and above the microwave range, above the maximum frequency forwhich signal amplification would be possible for the FET contained inthe cross-section of the TWFET.

It is another object of the present invention to increase the signalpower amplification capability in a TWFET, operated at frequencies inthe microwave range and above the microwave range, above the signalpower amplification capability of the FET contained in the cross-sectionof the TWFET.

It is a further object of the present invention to provide signal powergain in a TWFET, operated at frequencies in the microwave range andabove the microwave range, above the maximum signal power gain thatwould be possible for the FET contained in the cross-section of theTWFET.

It is yet another object of the present invention to provide forparallel interconnections of the inventive TWFET's for handling signalsof relatively higher power.

These and other objects of the present invention are attained by forminga depletion region beneath a gate electrode wherein, in a planetransverse to the direction of signal propagation, a depletion regionedge has a first end portion located between the gate electrode and adrain electrode and a second end portion located between the gateelectrode and a source electrode; and separating the depletion regionedge from the drain electrode. Further improvements in the operation ofthe TWFET include adjusting the first end portion of the depletionregion edge to be closer to the gate electrode relative to the distancebetween the second end portion of the depletion region edge and the gateelectrode, controlling an effective conductivity of a semiconductor ofthe traveling-wave field effect transistor, and setting the length ofthe gate electrode at substantially 1.0

BRIEF DESCRIPTION OF THE DRAWING

For a better understanding of these and other objects of the presentinvention, reference is made to the detailed description of theinvention which is to be read in conjunction with the following figuresof the drawing, wherein:

FIG. 1 is a schematic cross-section of a prior art FET.

FIG. 2 is a schematic cross-section of an FET that embodies the presentinvention.

FIG. 3A is a top-view of a prior art TWFET in a coplanar transmissionline.

FIG. 3B is a block diagram of a parallel TWFET structure.

FIG. 4 is a schematic cross-section of an FET that embodies the presentinvention and depicts various parameters of the FET.

FIG. 5 is a schematic cross-section of an FET that embodies the presentinvention and depicts various parameters of the FET's depletion region.

FIGS. 6-8 are contour plots of donor density for an FET that embodiesthe present invention.

FIGS. 9-12 are contour plots of DC electron concentration for an FETthat embodies the present invention.

FIGS. 13-36 show the variation in the AC admittance matrix for specificFETs of the present invention in the "forward" configuration.

FIGS. 37-60 show the variation in the AC charge matrix for specific FETsof the present invention in the "forward" configuration.

FIGS. 61-72 show the variation in the AC admittance matrix for specificFETs of the present invention in the "reverse" configuration.

FIGS. 73-84 show the variation in the AC charge matrix for specific FETsof the present invention in the "reverse" configuration.

DESCRIPTION OF THE PREFERRED EMBODIMENT

The present invention makes use of a physical effect in the structure ofa TWFET called non-reciprocal inductive coupling. The description of thepreferred embodiment begins the discussion of non-reciprocal inductivecoupling with a description of ideal coupled transmission lines. Thisincludes a description of the inductance and admittance couplingmatrices in ideal coupled transmission lines, using the TEM modeanalysis. The description of the preferred embodiment then uses the TEMmode analysis to discuss the operation of the TWFET structure, in whichthe inductance and admittance coupling matrices can be non-reciprocal.The TEM mode analysis provides a good approximation to the distributionof transverse electric and magnetic fields of the propagating mode inthe TWFET. These transverse electric and magnetic fields are theequivalent, in the TWFET structure, to the voltage and current waves ofthe transmission line.

In the case of ideal transmission lines, perfectly conducting rods areused to guide the propagating wave. These rods are imbedded in ahomogeneous, linear medium, are aligned along the direction of wavepropagation, and are uniform along this axis. A transmission linestructure of n coupled transmission lines is formed from (n+1)conducting rods. The TEM mode which propagates along this transmissionline structure can be considered to provide a voltage wave in the planeorthogonal to the direction of propagation. This voltage wave arisesfrom the nature of the TEM mode which allows an electrostatic potentialto be defined in the plane orthogonal to the direction of propagation.Consequently, in this transverse plane, each of the n conductors canhave a potential defined relative to a reference potential on the(n+1)rst conductor. This (n+1)rst conductor is often designated as theground conductor of the coupled transmission lines.

Accompanying this voltage configuration within this plane is adistribution of charge quantities on the (n+1) conductors. This chargedistribution propagates with the voltage wave, varying with time andalong the direction of propagation, to provide the current wave of theTEM mode. In the case of a single transmission line formed from twoconducting rods, the voltage and current distributions are what iscommonly known as the voltage and current of the transmission line. In asystem of n coupled transmission lines, any propagating TEM wave can beformed from a superposition of the eigenmodes of voltage and currentwaves for the system of transmission lines.

The equation for the propagation of the voltage wave of an eigenmode forthe system of transmission lines provides an important relationshipbetween the charge coupling and the inductive coupling of thetransmission lines.

The voltage eigenmode is defined as:

    v=v.sub.o e.sup.(jwt-yz)

where v is a vector of the voltages on the n conductors relative to the(n+1)rst reference conductor, j is √-1, w is the angular frequency, z isthe direction of propagation, t is time, and y is the propagationconstant of the voltage wave eigenmode.

The charge matrix, K, which relates the voltage and charge distributionfor the n transmission lines, is defined as:

    q=Kv

where q is a vector of charges on the n conductors, and K is an n×nmatrix relating the charge vector, q, to the voltage vector v.

Based on the preceding notation, the relationship between the voltageand current eigenmodes is defined as: ##EQU1## where v and i representvoltage and current vectors, and ε and μ_(o) represent the dielectricconstant and permeability of the medium surrounding the transmissionlines. In this equation, the permeability of the surrounding medium isassumed to be the permeability of free space. Therefore, μ=μ_(o) isused. This result can be recognized as the transmission line equationfor the z--derivative of the voltage eigenvector. It can be seen tofollow from the relationship between the inductive coupling matrix, Land the charge matrix, K: ##EQU2## where v_(c) is the speed of light inthe medium containing the transmission lines. In this case of idealtransmission lines, the impedance coupling matrix, Z, is equated to theproduct jωL.

The preceding relationship between voltage and current vectors providesone of the two well-known transmission line equations. The otherequation is defined as: ##EQU3##

This last equation implies a definition for the admittance matrix of thecoupled transmission lines which can be stated as: ##EQU4##

In these equations, σ is the conductivity of the surrounding medium, andε is its dielectric constant

In this case of ideal transmission lines, the Y matrix can be seen todescribe the transverse current flow between the electrodes of thetransmission lines with the presence of an inter-electrode potential inthis transverse plane. For example, consider the case of idealtransmission lines imbedded in a homogeneous medium with finiteconductivity. The Y matrix would describe the shunt conductance of anequivalent parallel R-C circuit element model for a differential lengthsection of the structure, with conductance given by the term ##EQU5##and capacitive coupling of these lines given by the term

    (jωK)

The Z matrix, which is contained in the other transmission line equationfor the z-derivative of the voltage eigenvector, describes the change inthe voltage with longitudinal direction as related to current flow alongthe electrodes in the direction of signal propagation in the coupledtransmission lines. The i vector corresponds to the surface current flowalong the electrodes, which arises with the presence of surface chargeson these conductors. In view of this fact, it can be seen that Z, L, andK matrices all pertain to the longitudinal current flow, which ispresent in all transmission line structures.

In this case of ideal transmission lines (assuming the surroundingmedium is linear and reciprocal), it is important to note that thecharge matrix of the ideal (passive) transmission line structure isreciprocal and provides reciprocal inductive coupling in thesestructures. This can be defined in terms of a matrix element, m_(kp) inwhich m_(kl) =m_(lk). This means that the charge induced on conductor lby the presence of a voltage on conductor k is equal to the chargeinduced on conductor k by the presence of the same voltage on conductorl.

The equivalences between voltage and current on a transmission line totransverse electric and magnetic fields in non-ideal transmission linestructures allows the use of the same equations for ideal coupledtransmission lines for the analysis of the TWFET. A detailed derivationof these transmission line equations is presented in Appendix A. Inaddition, the TEM mode analysis provides a good approximation to thetransverse electric and magnetic fields. Therefore, it will be used hereto describe the operation of the present invention.

In the present TWFET invention, the Y and Z matrices are considered as2×2 matrices for the 3 conductors formed by the elongated source, gate,and drain electrodes. While a common source configuration is assumed, itwill be obvious to those skilled in the art that the analysis caninclude other configurations.

In a TWFET, the Y matrix elements are complex valued and describe theFET activity in the plane transverse to the direction of signalpropagation by relating the inter-electrode transverse current flow toAC potential difference between the electrodes. Most often, thisactivity is evaluated using circuit models. An important difference ofthe Y matrix of the TWFET structure as compared to that of the idealtransmission lines is the nonreciprocity in the Y matrix of the TWFET.At low frequencies, this property appears in the transconductancecomponent of circuit element models for the FET. The presence of thistransconductance provides non-reciprocity in that the AC gate voltagetransfers an AC particle current to the drain electrode, but an AC drainvoltage does not transfer an approximately equal AC particle current tothe gate electrode. This is an approximate description of thenon-reciprocity feature of the Y matrix of the conventional FET which ismade use of in both the present invention and prior art TWFETstructures.

The important difference in the present invention, as compared to thatof the prior art, is in the design of the properties of the Z matrix.This design difference is found in the inductive coupling matrix whichprovides part of the Z matrix, with the series resistance on the gateand drain electrodes providing the remaining component:

    Z=R+jωL

In the case of the 3 conductor system of the source, gate, and drainelectrodes, R is a 2×2 diagonal matrix containing the gate and drainelectrode series resistances, and L is the 2×2 inductive coupling matrixfor the TWFET. In the present invention, the inductive coupling matrixis designed to have a significant non-reciprocal property.

In the prior art, the inductive coupling matrices have been left as anunspecified component of the TWFET design, or have been analyzed ashaving reciprocal properties. This reciprocal inductive coupling hasbeen obtained through calculations of the passive electrodes of theTWFET, or has also been obtained with calculations that includecapacitances of circuit models for the active FET structure.

The prior art FIG. 1 is used to explain how this reciprocal inductivecoupling is obtained with these calculations. In prior art FIG. 1, theFET 120 formed in the cross-section of a TWFET in the plane transverseto the direction of signal propagation comprises a semiconductormaterial 104, a source electrode 100, a drain electrode 102 and a gateelectrode 106. The gate electrode 106 can be either metallic--in whichcase it forms a Schottky contact to the semiconductor material 104, orit can be formed as a semiconductor of opposite type to thesemiconductor material 104. When the semiconductor material 104 is ann-type material and the gate electrode 106 is formed as a p-typesemiconductor, the gate electrode 106 is reverse-biased--forming arectifying contact to the semiconductor material 104 and changing theFET 120 to a junction FET (JFET). In either configuration, eithermetallic or formed as a semiconductor, the gate electrode 106 isreverse-biased so that little or no DC current flows into thesemiconductor material 104 and so that a depletion region 108 is formedunder the gate electrode 106.

In prior art FIG. 1, a source electrode region 112 and a drain electroderegion 114 can be metals making ohmic contact with the semiconductormaterial 104 or formed with semiconductor doping. For example, whensemiconductor material 104 is n-type, the source electrode region 112and the drain electrode region 114 are highly doped n⁺ regions--formingquasi-metallic ohmic regions. These n⁺ -ohmic regions provide anextension to the metallic conductors 116 that contact the sourceelectrode region 112 and the drain electrode region 114. A depletionregion edge 110 is in contact with the source electrode region 112 ofthe source electrode 100 and the drain electrode region 114 of the drainelectrode 102.

The calculation of the inductive coupling matrix for this prior artstructure proceeds from the calculation of the K matrix, as describedearlier for the case of the ideal transmission lines. In thiscalculation, two AC-bias cases are considered: (a) An AC bias, v_(o),applied to the gate electrode 106 with the source electrode 100 and thedrain electrode 102 maintained at an AC bias of 0 volts; and (b) An ACbias, v_(o), applied to the drain electrode 102 with the sourceelectrode 100 and the gate electrode 106 maintained at an AC bias of 0volts. For the common source configuration, an AC bias, v_(o), refers toan AC excitation of complex value maintained between one electroderelative to the source electrode 100, while an AC bias of 0 volts meansthat the electrode is kept at the same AC potential as the sourceelectrode 100.

In case (a), the charge induced on the gate electrode 106 and the drainelectrode 102 is respectively:

    q.sub.g,a=K.sub.g-s, v.sub.o +K.sub.g-d O

    q.sub.d,a =K.sub.d-g v.sub.o +K.sub.d-s O

In case (b), the charge induced on the gate electrode 106 and the drainelectrode 102 is respectively:

    q.sub.g,b =K.sub.g-s O+K.sub.g-d v.sub.o

    q.sub.d,b=K.sub.d-g O+K.sub.d-s v.sub.o

In these equations, q_(g),a refers to the AC charge on the gateelectrode 106 with AC bias case(a), q_(d),a refers to the AC charge onthe drain electrode 102 with AC bias case(a), q_(g),b refers to the ACcharge on the gate electrode 106 with AC bias case(b), and q_(d),brefers to the AC charge on the drain electrode 102 with AC bias case(b). K_(g-s), K_(g-d), K_(d-s), and K_(d-g) are the elements of thecharge matrix, K. The elements of the first column of the K matrix areK_(g-s) and K_(d-g),. These correspond respectively to the ratio of thecharge induced on the gate electrode 106 and the drain electrode 102 tothe voltage applied to the gate electrode 106 with AC-bias case (a). Theelements of the second column of the K matrix are K_(g-d) and K_(d-s).These correspond respectively to the ratio of the charge induced on thegate electrode 106 and the drain electrode 102 to the voltage applied tothe drain electrode 102 with AC-bias case(b). The off-diagonal elementsof the K matrix are K_(d-g) and K_(g-d).

As prior art FIG. 1 suggests, the interaction of the gate electrode 106and the drain electrode 102 in these two cases is well-approximated by asimple capacitive coupling between the electrodes. In particular, thecharge induced on the drain electrode 102 with the AC bias applied tothe gate electrode 106 (case(a)) is approximately equal to the chargeinduced on the gate electrode 106 with the AC bias applied to the drainelectrode 102 (case(b)). This shows that the off-diagonal elements ofthe K matrix, K_(g-d) and K_(d-g), are approximately equal to eachother. As a result, the K, L, and Z matrices are approximatelyreciprocal.

The preceding analysis of the K matrix calculation shows that the simplecapacitive coupling of the gate electrode 106 and the drain electrode102 of the prior art TWFET structure leads to reciprocal charge transferand reciprocal K, L, and Z matrices. In a similar manner, when the Kmatrix is calculated using capacitances taken from a circuit model forthe active FET, while omitting series resistance components of thiscircuit model, the charge transfer is also reciprocal, as are the K, Land Z matrices resulting from this analysis.

In FIG. 2, the FET 5 formed in the cross-section of a TWFET in the planetransverse to the direction of signal propagation comprises asemiconductor material 14, a source electrode 10, a drain electrode 12and a gate electrode 16. The gate electrode 16 can be eithermetallic--in which case it forms a Schottky contact to the semiconductormaterial 14, or it can be formed as a semiconductor of opposite type tothe semiconductor material 14. In either configuration, either metallicor formed as a semiconductor, the gate electrode 16 is reverse-biasedherein defined as meaning a bias so that little or no DC current flowsinto the semiconductor material 14, and so that a depletion region 18 isformed under the gate electrode 16.

In the same manner as the gate electrode 16, a source electrode region22 and a drain electrode region 24 can be metals making ohmic contactwith the semiconductor material 14 or formed with semiconductor doping.For example, when semiconductor material 14 is n-type, the sourceelectrode region 22 and the drain electrode region 24 are highly dopedn⁺ regions--forming quasi-metallic ohmic regions. These n⁺ -ohmicregions provide an extension to the metallic conductors 2 that contactthe source electrode region 22 and the drain electrode region 24. InFIG. 2, a depletion region edge 20 is in contact with the sourceelectrode region 22 of the source electrode 10, but it need not makecontact with the source electrode region 22. However, as shown in FIG.2, the depletion edge 20 cannot make contact with the drain electroderegion 24.

An AC potential difference applied between the gate electrode 16 and thedrain electrode 12 is distributed across the depletion region 18 and aneutral region 26 in the plane of FIG. 2. This part of the FET 5 can beconsidered to act as an R-C series circuit. The gate electrode 16 andthe depletion region edge 20 form a capacitor 30, while the neutralregion 26 provides a small series resistance 32 between the depletionregion edge 20 and the drain electrode 12. The conductivity of thesemiconductor material 14, and the distance of separation of thedepletion region edge 20 and the drain electrode 12, can be selected sothat most of the AC voltage which appears between the gate electrode 16and the drain electrode 12 will appear across the depletion region 18,represented by the capacitor 30, and a comparatively small voltage willappear across the neutral region 26, represented by the resistor 32. Inthis case, almost all of the AC charge which appears on the gateelectrode 16 is accompanied by changes in the depth of the depletionregion 18. The comparatively small amount of AC voltage which isdistributed across the neutral region 26 causes a correspondingly smallAC charge to appear on the surface of the drain electrode 12.

In calculating the K matrix, the preceding argument shows that thecharge induced on the drain electrode 12 with the AC bias of case(a)will be significantly less than the charge induced on the gate electrode16 with the AC bias of case(b). As a result, the off-diagonal elementsof the K matrix, K_(g-d) and K_(d-g), have significantly differentvalues. Consequently, in the present invention, the K, L, and Z matricesare significantly non-reciprocal. In the description of the presentinvention, this effect shall be referred to as non-reciprocal inductivecoupling.

The present invention obtains non-reciprocal inductive coupling by meansof the separation of the depletion region edge 20 and the drainelectrode 12. The separation is created by maintaining a region ofsemiconductor material 28 which provides a large enough distance betweenthe depletion region edge 20 and the drain electrode 12, such that thedepletion region edge 20 can screen the electric field from the drainelectrode 12, as described above. The region of semiconductor material28 between the depletion region edge 20 and the drain electrode 12should also have sufficiently low conductivity--to establish adistinction between the region of semiconductor material 28 and thedrain electrode region 24 of the drain electrode 12. As this separationbetween the depletion region edge 20 and the drain electrode 12 isdecreased, or as the conductivity of the region of semiconductormaterial 28 is increased, the limit of reciprocal inductive coupling isobtained.

The preceding description of the present invention has used a TEM modeanalysis to describe non-reciprocal inductive coupling by means ofnon-reciprocal charge transfer. As noted earlier, the TEM mode analysisprovides an approximate description for transverse electric and magneticfields in the structure of a TWFET.

The surface charge on ideal transmission line electrodes is proportionalto the tangential magnetic fields at the surface of these conductors.Consequently, the non-reciprocal charge transfer of the preceding TEMmode analysis of FIG. 2 corresponds to non-reciprocity in the transversemagnetic fields at the surface of the gate electrode 16 or drainelectode 12. The non-reciprocal inductive coupling of the presentinvention can thus be understood in terms of non-reciprocity in thecharge transfer and associated non-reciprocity in the K matrix, or as anequivalence in non-reciprocity of the transverse magnetic fields whichalso correspond to non-reciprocity in the L and Z matrices due to thenon-reciprocity of the longitudinal currents.

An importance consequence of the design of TWFETs with non-reciprocalinductive coupling is the increase in the signal power gain in thetransistor. This is shown in the following pair of differentialequations governing the TWFET structure: ##EQU6##

In these equations, Z is defined as the sum of the diagonal seriesresistance matrix, R and the product of jω with L, the inductivecoupling matrix. As noted earlier, R allows for series resistance on thetransmission line electrodes. The parallel roles of Y and Z in theseequations indicate that non-reciprocity in the inductance matrix can bejust as important as non-reciprocity in the admittance matrix inproviding power gain for signal amplification in the TWFET.

An important special consequence of the use of non-reciprocal inductivecoupling in TWFETs is that it can increase the range of frequency ofsignal gain in the TWFET above that which is possible when only theeffect of the non-reciprocal admittance matrix is available, such as inprevious TWFET designs. In conventional transistors, i.e. those notdesigned as a part of an active coupled transmission line structure,there is a maximum frequency of operation above which the transistorcannot be used to provide gain to signals. In this discussion, thisfrequency limit is referred to as f_(MAX) for the transistor. In thecase of the conventional FET which forms the cross-section of the TWFET,f_(MAX) represents the upper frequency limit for obtaining signal gainfrom the Y matrix in the TWFET. When non-reciprocal inductive couplingis present, TWFETs can provide signal power gain at frequencies abovef_(MAX) of the conventional FET which lies in the plane transverse tothe direction of signal propagation in the TWFET. The power gain atfrequencies above f_(MAX) can occur when the Z coupling matrix exhibitsthe mathematical property that z_(oh), the hermitian part of Z, definedas Z_(oh) =1/2(Z+Z*), (where Z* denotes the complex conjugate transposeof Z), is not non-negative definite, which is to say that Z_(oh)satisfies x*Z_(oh) x<0 for some nonzero complex valued vector, x, withcomplex conjugate transpose x*. It should be noted that this is aminimal requirement for power gain in the TWFET. As will be obvious tothose of ordinary skill in the art, the success of the design depends onthe relative sizes of matrix elements in the Z and Y coupling matrices,in addition to this minimal requirement that the hermitian part ofeither Y or Z is not non-negative definite as defined above. A detailedanalysis of the conditions which allow an increase in average power flowalong a pair of coupled transmission lines is presented in Appendix B.

While non-reciprocal inductive coupling has been discussed in the TWFETstructure, it can be obtained in similar distributed circuit elementsbased on the combination of an active semiconductor transistor withcoupled transmission lines.

The essential requirement is that the cross-section of the distributedcircuit element sustain AC surface charges on the electrodes or havetransverse magnetic fields near the electrode surfaces which exhibitnon-reciprocity as described above.

In the present invention, the TWFET designs are based on the use of adual-FET structure, as disclosed in U.S. Pat. No. 4,587,541. Thedual-FET structure allows the design of a single conventional FET withthe understanding that the symmetry of the propagating signal in thisstructure has the effect of doubling the contact area of the electrodes.This doubled area implies that transforming the results of a single FETto those of a symmetric dual-FET requires doubling the calculated valuesfor the admittance and charge matrix elements of a single FET andreducing the electrode series resistance values by a factor of 2.

The symmetry of the dual-FET allows the designs of the present inventionto be developed with a 2-dimensional simulation of a single FETstructure. This simulation is based on the finite element solution ofthe standard semiconductor equations using a TEM analysis of the TWFET.The semiconductor equations used for the present invention are those ofan n-type semiconductor, neglecting generation and recombinationeffects: ##EQU7## In these equations, IqI represents the magnitude ofthe electronic charge, ε represents the dielectric constant, nrepresents the electron concentration, N_(d) represents the donorconcentration, and φ represents the electrostatic potential, as would beconsistent with a TEM analysis of the TWFET. In addition, J_(n)represents the electron current, with an electron mobility of μ_(n), andan electron diffusion coefficient of D_(n).

The simulation calculates the small-signal AC semiconductor devicebehavior in accordance with a method which is almost identical to thesinusoidal steady state analysis method in S. E. Laux, "Techniques forSmall-Signal Analysis of Semiconductor Devices," IEEE Transactions onElectron Devices Vol. ED-32(10), pp. 2028-2037 (October, 1985).Following the method described earlier for the AC charge matrixcalculations, two AC bias cases are used to calculate the total ACsurface charge on the gate electrode 16 and the drain electrode 12 ofFIG. 2, and to calculate the total AC current flowing into theseelectrodes. The AC surface charge on the electrodes is calculated fromthe normal derivative of the AC potential at these electrode surfaces Ineach AC bias case, the AC voltage is applied to one electrode relativeto the remaining two electrodes at an AC voltage of 0. Thesecalculations yield admittance and charge matrices by the method whichwas described in detail earlier for the charge matrix calculation.

In this sinusoidal steady state analysis, the complex phasor notation isused so that the time derivative in the semiconductor equations isreplaced by a factor of (jω), and complex valued admittance and chargematrices are obtained. The results for admittance and charge matricesare converted to the inductance and admittance coupling matrices of thedistributed dual-FET using the formulas given earlier for obtaining theinductance matrix from the charge matrix and the use of the factor oftwo scaling noted above. The coupling matrices are combined withselected values of series resistance for the dual-FET electrodes. Inthis way, the use of the 2-dimensional semiconductor device simulationof a fully specified FET in combination with series resistance valuesyields the complete coupling matrix information for the dual-FETstructure.

In FIG. 3A, the TWFET is designed with open circuit terminations at theends of the gate electrodes 16, 16' and the drain electrode 12, whichare opposite to the attachments to first coplanar transmission lines 34or second coplanar transmission lines 36. A 2-port impedance andadmittance matrix can be calculated for the TWFET of FIG. 3A as afunction of the coupling length, z_(o), using an extension of the methodof V. K. Tripathi, "Asymmetric Coupled Transmission Lines in anInhomogeneous Medium," IEEE Trans. Microwave Theory Tech. Vol.MTT-23(9), pp. 734-739 (September, 1975) to treat the case of general(non-reciprocal) Y and Z TWFET coupling matrices. In the presentinvention, the 2-port admittance matrix was calculated for the couplinglengths, z_(o), ranging from 5×10⁻⁴ cm to 5×10⁻² cm. This admittancematrix data allows the calculation of maximum unilateral gain (U,Mason's U- function), maximum available gain (MAG), maximum stable gain(MSG), and the stability parameter (k) for the TWFET, using the standardformulas for these quantities as a function of the admittance orimpedance matrix elements.

In the present invention, the electrode series resistance is calculatedfrom the skin depth of the conductors. The resistance-per-unit length,which can be defined as the ratio of resistivity to cross-sectionalarea, uses the cross-sectional area formed by the product of skin depthand electrode width (or circumference). With this definition, a metallicresistivity of 10⁻⁶ ohm-cm combined with an electrode width of 10⁻⁴ cmand a skin depth of 10⁻⁵ cm provides a series resistance of 10³ ohm/cmor 1 k-ohm/cm. When the dual-FET structure is used, this seriesresistance value would reduce to 500 ohm/cm.

It should be noted that the capacitive effects due to the abrupttermination of the end of the gate electrode and drain electrode, andthe inter-electrode capacitive coupling from the transverse electricfields above the surface of the semiconductor have been omitted from thepresent analysis. A person of ordinary skill in the art would understandthat the corrections for these effects would be small.

In FIG. 3B, four of the TWFET structures of FIG. 3A are arranged in aparallel configuration for handling signals with relatively higherpower. In other preferred embodiments two, three or more than fourTWFET's may be parallelled. In FIG. 3B, a single transmission line 11carries an input signal of a relatively high power, compared to thepower handling of any one of the TWFET's 17. The input signal on thesingle transmission line 11 is divided by a network 13 into four signalswhich are substantially identical in power and phase, each signal havingabout one fourth the input signal power. Each of these four signals arefed via identical transmission lines 15 to the input side of fourTWFET's 17 each of which can easily handle the divided power levels.

The output of each of the TWFET's 19 can be used independently or any ofthe outputs 19 may be combined to form a higher powered output signal.Alternatively, the four TWFET's, in fact, operate in parallel when theinput signals are in-phase no matter what the input signal source maybe. In fact, in other preferred embodiments, the network 13 and/or 13'may not be used. As known in the art, when the TWFET's are used forsignal amplification, for optimal power transfer, the input and outputimpedances of the surrounding circuitry should present a conjugate matchto the TWFET input and output. In this preferred embodiment, impedancematching circuitry (not shown) could be designed for location on thesingle transmission line 11. Alternatively, impedance matching networks(not shown) may be designed for location at the transmission lines 15which connect to the inputs of the four TWFET's 17.

As referenced above and as known in the art, the outputs 19 of the fourTWFET's may be connected to transmission lines that lead to a networksimilar to network 13. In this case the network 13' receives the signalscarried on the transmission lines 19' from the four TWFET outputs andcombines them into a signal of relatively higher power on the singletransmission line 21. Impedance matching circuitry could be designed forlocation at the single output transmission line 21 or, alternatively,impedance matching circuitry could be designed for location at thetransmission lines 19'. The transmission lines 15 and 19' may bemicrostrip or coplanar transmission lines. When coplanar transmissionlines are used, the TWFET's may be of the type shown in FIG. 3A.

The power dividing networks 13 and 13' may use conductive lines formedby the branching of a single transmission line conductor to formmultiple identical transmission line conductors. When such branching isused, one of the branched transmission line conductors may cross overanother using a structure known as an air bridge or using otherstructures which use dielectric materials to insulate the metallicelectrodes when one metal electrode crosses over another.

Directional couplers formed with coupled transmission lines in which thecoupled transmission line structure is designed to function as a powersplitter may also be used in-power dividing networks. These powersplitters may be combined in series so that a signal from a singletransmission line is fed into a power splitter which in turn transfersthe signal into two more power splitters, each of which transfers thesignal into two more output transmission lines. This combination ofthree power splitters would create a power dividing network which cantransfer an incoming signal on a single transmission line to outputsignals on four transmission lines. This concept may be repeated withcombinations of more power splitters to further divide the power of theinput signal as needed in various embodiments of the present invention.

In the power dividing stuctures, care should be taken to provideidentical signals, having identical phases to the input of each TWFET.

In FIG. 4, S_(SG) represents the source-to-gate inter-electrodeseparation distance, S_(GD) represents the gate-to-drain inter-electrodeseparation distance, and a represents the depth of a channel 38.

The n-type material used for the FET semiconductor has a conductivity ofapproximately 1.6 Siemens/cm for most of the semiconductor region. Forexample, an n-type semiconductor with a donor concentration of 10¹⁶ cm⁻³and a constant electron mobility of 10³ cm² /V-sec could be used. Thisconductivity could be obtained with n-type GaAs composed of a net donordensity to obtain this equilibrium electron concentration, but with anelectron mobility which is reduced by means of some partial compensationof the n-type material with p-type acceptors or by some damage to thematerial. Those of ordinary skill in the art will recognize that whenn-type silicon is used little or no modification to the donor density orelectron mobility is needed. Since actual electron mobility insemiconductor devices is a function of many physical parameters,including the electric field strength, the specified conductivity of 1.6mhos/cm can be obtained by locally adjusting the net donorconcentration, as needed.

In addition to the TWFET examples with uniform conductivity and uniformnet donor density of 10¹⁶ cm⁻³, three examples with local regions ofincreased doping density (and correspondingly increased conductivity)were used for the present invention. These TWFET structures withnon-uniform doping density used distributions of net donorconcentrations as shown in FIGS. 6 through 8. FIGS. 6 through 8 show thecontours of net donor concentrations in the FET in the plane transverseto the direction of signal propagation in the TWFET. In FIG. 6, asolid-line contour 40 has a value of 1.5×10¹⁶ cm⁻³ and a dashed-linecontour 42 has a value of 3.5×10¹⁶ cm⁻³. The contour lines 40 and 42indicate the local region of increased donor density with the rest ofthe semiconductor of the FET having a background donor concentration of10¹⁶ cm⁻³. The maximum value of the net donor concentration in the localregion of increased donor density is approximately 3.9×10¹⁶ cm⁻³,located within the dashed-line contour 42.

In FIG. 7, a similar donor density contour plot for a different donordensity distribution is shown. As in the case of FIG. 6, FIG. 7indicates a local region of increased net donor density with the rest ofthe semiconductor of the FET having a background donor concentration of10¹⁶ cm⁻³. In FIG. 7, a solid-line contour 44 has a value of 1.5×10¹⁶cm⁻³ and a dashed-line contour 46 has a value of 4.0×10¹⁶ cm⁻³. Themaximum value of the net donor concentration in the local region ofincreased donor density is approximately 5.2×10¹⁶ cm⁻³, located withinthe dashed-line contour 46.

In FIG. 8, another similar donor density contour plot for a third donordensity distribution is shown. As in the cases of FIG. 6 and FIG. 7,FIG. 8 indicates a local region of increased net donor density with therest of the semiconductor having a background donor concentration of10¹⁶ cm⁻³. In FIG. 8, a solid-line contour 54 has a value of 1.5×10¹⁶cm⁻³ and a dashed-line contour 56 has a value of 4.0×10¹⁶ cm⁻³. Themaximum net donor concentration in the region of increased donor densityis approximately 4.2×10¹⁶ cm⁻³, located within the dashed-line contour56. These local regions of increased net donor density, shown in FIGS.6-8, can be created using typical device fabrication techniques. Forexample, the local regions can be formed with ion implantation through amask window on the surface of a semiconductor of uniform donor densityof 10¹⁶ cm⁻³.

In the present invention, the Schottky barrier height used for the TWFETstructures was 0.7505 eV. This barrier height value corresponds to thevalue of built-in potential obtained with the abrupt depletion model ofthe junction of the Schottky contact with the n-type semiconductor. As aresult, it specifies a built-in potential for the Schottky contact tothe semiconductor, not a true barrier height value.

In FIG. 5, the general features of the depletion region 18, as used inthe TWFETs of the present invention, is shown. In the present invention,the shape of the depletion region 18 is a mirror-image of the shape of adepletion region found in the prior art FETs. The shape of the depletionregion 18 is obtained when the depletion region edge 20 is further awayfrom the gate electrode 16 at the end near the source electrode 10 thanthe depletion region edge 20 is from the gate electrode 16 at the endnear the drain electrode 12. The deepest point of the depletion region18 is located almost directly below the end of the gate electrode 16near the source electrode 10. The parameter X_(SG) represents thehorizontal distance from the depletion region edge 20 to the gateelectrode 16 between the source electrode 10 and the gate electrode 16,X_(GD) represents the horizontal distance from the gate electrode 16 tothe depletion region edge 20 between the gate electrode 16 and the drainelectrode 12, IXI_(max-depth) represents the absolute value of thedifference in horizontal distance from the deepest point of thedepletion region 18 as compared to the location of the end of the gateelectrode 16 nearest the source electrode 10, and Y_(max-depth)represents the vertical distance at the deepest point in the depletionregion 18.

In the abrupt depletion model of the FET, the depletion region edge 20can be considered as the boundary between the neutral region 26 inwhich, for n-type semiconductors, the electron concentration isapproximately equal to the doping density, and the depletion region 18,in which the electron concentration is approximately zero. In thepresent invention, the depletion region edge 20 is defined as the pointat which the electron concentration decreases to half of the donorconcentration of the majority of the semiconductor in the FET. In thepresent invention, this is a concentration value of 5×10¹⁵ cm⁻³.

The shape of the depletion region 18 has been obtained in the presentinvention by use of negative DC-bias on the drain electrode 12 relativeto the source electrode 10. The gate electrode 16 and the drainelectrode 12 are supplied with DC biasing by the use of bias tees (notshown) attached to the coplanar transmission lines 34 and 36. It ispossible that the DC-bias value of the drain electrode 12 can create atotal potential difference for the contact junction between the gateelectrode 16 and the drain electrode 12 which is almost zero or is quiteclose to a forward bias condition for the gate electrode 16 relative tothe drain electrode 12. In order to avoid being this close to a forwardbias condition for the gate electrode 16, the doping density between thegate electrode 16 and the depletion region edge 20 near the drainelectrode 12 should be increased. This allows the same shape of thedepletion region 18 to be obtained with a larger reverse bias betweenthe gate electrode 16 and the drain electrode 12. The same technique canbe applied to obtain the same shape of the depletion region 18 with apositive DC-bias on the drain electrode 12 relative to the sourceelectrode

A series of TWFETs were designed, as discussed above, that provided gainabove the f_(MAX) of the FET contained in the cross-section of theTWFET. In the first series, referred to as the "forward" configuration,the signal was applied using the symmetric mode of operation forcoplanar transmission lines shown in FIG. 3A. Thus, the signal input wasapplied identically to the pair of electrodes of the source electrode 10and the gate electrode 16, and to the pair of electrodes of the sourceelectrode 10' and the gate electrode 16'. The signal output was takenfrom the pair of electrodes of the source electrode 10 and the drainelectrode 12, and the pair of electrodes of the source electrode 10' andthe drain electrode 12. Table I shows the gain values for this series of"forward" configuration TWFETs.

                  TABLE I(a)    ______________________________________                                       Gate  Donor           V.sub.gate                   V.sub.drain                           S.sub.SG                                 S.sub.GD                                       Length                                             Distribution    CASE   volts   volts   microns                                 microns                                       microns                                             Type    ______________________________________    1      -.4988  -1.103  0.5   1.0   1.0   Uniform    2      -.4988  -1.203  1.0   0.5   1.0   Uniform    3      -.4988  -1.203  1.0   0.5   0.25  Uniform    4      -.4988  -1.203  0.5   0.5   0.25  Uniform    5      -.4988  -1.203  1.0   0.5   1.0   FIG. 6    6      -.4988  -1.203  1.0   0.5   1.0   FIG. 8    ______________________________________

The DC-bias values are relative to the source electrode 10. The sourceelectrode 10 is DC-biased at 0 V. In Cases 1-4, the donor distributionhas a uniform concentration of 10¹⁶ cm⁻³. In Case 5, the donordistribution has a local region of increased concentration--as shown inFIG. 6. In Case 6, the donor distribution has a local region increasedconcentration--as shown in FIG. 8.

                  TABLE I(b)    ______________________________________           X.sub.SG                   X.sub.GD                           |X|.sub.max-depth                                  Y.sub.max-depth                                         Depletion Region    CASE   microns microns microns                                  microns                                         Type    ______________________________________    1      0.3     0.1     0.1    0.3    FIG. 9    2      0.3     0.1     0.1    0.3    FIG. 9    3      0.3     0.1     <0.1   0.2    FIG. 9    4      0.3     0.1     <0.1   0.2    FIG. 9    5      0.3     0.1     0.1    0.3    FIG. 10    6      0.3     <0.1    <0.1   0.2    FIG. 12    ______________________________________

In Cases 1-4, the shape of the depletion region 18 is of the shape shownin the DC electron concentration contour plot of FIG. 9. In FIG. 9, asolid-line electron concentration contour 48 has a value of 5×10¹⁵ cm⁻³,the value assigned to the depletion region edge as earlier noted. InCase 5, the shape of the depletion region 18 is of the shape shown inthe DC electron concentration contour plot of FIG 10. In FIG. 10, asolid-line electron concentration contour 50 has a value of 5×10¹⁵ cm⁻³,the value assigned to the depletion region edge as earlier noted. InCase 6, the shape of the depletion region 18 is of the shape shown inthe DC electron concentration contour plot of FIG. 12. In FIG. 12, asolid-line electron concentration contour 58 has a value of 5×10¹⁵ cm⁻³,the value assigned to the depletion region edge as earlier noted.

The values for X_(sg), X_(gd), IXI_(max-depth), and Y_(max-depth) arerounded to the nearest 0.1 micron.

                  TABLE I(c)    ______________________________________                 a      R.sub.g                             R.sub.d                                  U      MAG    MAG          f.sub.MAX                 mi-    ohm/ ohm/ (Z.sub.0,U)                                         (Z.sub.0,MAG)                                                (Z.sub.0,MAG)    CASE  GHz    crons  cm   cm   (10.sup.-2 cm)                                         (10.sup.-2 cm)                                                (10.sup.-2 cm)    ______________________________________    1         10     0.43 500  500  1.3    1.46   NA                                    (3.1)  (3.1)    2   (a)   10     0.43 625  625  4.6    2.0    3.45                                    (3.2)  (3.6)  (3.5)        (b)               750  750  1.2    1.26   NA                               (3.1)                                    (3.1)    3   (a)   15     0.43 625  625  NA     1.85   3.67                                           (3.6)  (3.4)        (b)               690  625  5.4    1.6    3.57                                    (3.1)  (3.5)  (3.4)        (c)               750  750  2.4    1.86   2.85                                    (3.0)  (3.3)  (3.2)    4   (a)   30     0.43 350  275  3.7    2.34   3.56                                    (3.1)  (3.4)  (3.3)        (b)               400  300  1.6    2.22   2.4    5                               (3.0)  (3.1)  (3.0)              <1.0   0.43 275  275  NA     1.8    2.54                                           (3.6)  (4.15)    6         <1.0   0.43 275  275  NA     3.1    4.6                                           (4.0)  (3.9)    ______________________________________

In Cases 1-4 and 6, the gain data are for a 100 GHz signal frequency. InCase 5, the gain data is for a 70 GHz signal frequency. In Cases 1-4,the gain values were observed over a range of coupling lengths of 2×10⁻³cm to 5.1×10⁻² cm. In Cases 5-6, the gain values were observed over arange of coupling lengths of 5×10⁻⁴ cm to 5×10⁻² cm. The columns of U,MAG, and MSG contain the maximum applicable values for these gainparameters which were observed for these ranges of coupling lengths. Asthose with ordinary skill in the art are aware, MAG is a valid gainparameter when the stability parameter is greater than unity, and MSG isa valid gain parameter when the stability parameter, k, is less thanunity, but positive. In Cases 1 and 2(b), the use of "NA" for MSG valuesindicates that no coupling lengths were observed to have k less thanunity, therefore MSG was not applicable to these cases. In Cases 3(a), 5and 6, the use of "NA" for U values indicates examples for which novalid maximum value of U could be assigned due to the appearance ofregions of the coupling length for which U was negative. Coupling lengthvalues which lie at the boundaries of these negative-U regions have aninfinite value for U. Consequently, in these cases, no maximum value forU can be assigned. In Table I(c), the coupling lengths and gain datashould be interpreted as approximate values and the f_(max) frequency isapproximated to within 5 GHz.

In FIGS. 13-36, the variation with frequency for the AC admittancematrix elements for the FETs in this series of "forward" configurationTWFETs is shown. FIGS. 13-16 show the AC admittance matrix elements forCase 1, FIGS. 17-20 show the AC admittance matrix elements for Case 2,FIGS. 21-24 show the AC admittance matrix elements for Case 3, FIGS.25-28 show the AC admittance matrix elements for Case 4, FIGS. 29-32show the AC admittance matrix elements for Case 5, and FIGS. 33-36 showthe AC admittance matrix elements for Case 6. The charge matrix elementvalues have been normalized to the value of the dielectric constant offree space, to yield a dimensionless quantity. In FIGS. 37-60, thevariation with frequency for the AC charge matrix elements for thisseries of "forward" configuration TWFETs is shown. FIGS. 37-40 show theAC charge matrix elements for Case 1, FIGS. 41-44 show the AC chargematrix elements for Case 2, FIGS. 45-48 show the AC charge matrixelements for Case 3, FIGS. 49-52 show the AC charge matrix elements forCase 4, FIGS. 53-56 show the AC charge matrix elements for Case 5, andFIGS. 57-60 show the AC charge matrix elements for Case 6. In FIGS.13-60 the notation used for the subscripts dg, ds, gd, and gs of thecharge and admittance matrix elements is explained in the detaileddescription of the charge matrix calculation. The curve markers indicatethe frequency values 1 GHz, 40 GHz, 70 GHz and 120 GHz in sequence foreach matrix element curve.

In the second series, referred to as the "reverse" configuration, thesignal was applied using the symmetric mode of operation for coplanartransmission lines shown in FIG. 3A. Thus, the signal input was appliedidentically to the pair of electrodes of the source electrode 10 and thedrain electrode 12, and to the pair of electrodes of the sourceelectrode 10' and the drain electrode 12. The signal output was takenfrom the pair of electrodes of the source electrode 10 and the gateelectrode 16, and the pair of electrodes of the source electrode 10' andthe gate electrode 16'. With this "reverse" configuration, the elementsof the coupling matrices and voltage and current vectors of thetransmission line equations must be re-ordered to correspond. Forexample, with this re-ordering of the matrix, the first column of the Zmatrix contains Z_(ds) and Z_(gd) and the second column of the Z matrixcontains Z_(dg) and Z_(gs), making the diagonal elements of the Z matrixZ_(ds) and Z_(gs). A similar re-ordering is also required for the Ymatrix. Table II shows the parameters and gain values for this series of"reverse" configuration TWFETs.

                  TABLE II(a)    ______________________________________                                       Gate  Donor           V.sub.gate                   V.sub.drain                           S.sub.SG                                 S.sub.GD                                       Length                                             Distribution    CASE   volts   volts   microns                                 microns                                       microns                                             Type    ______________________________________    1      -.4988  -1.203  1.0   0.5   1.0   Uniform    2      -.4988  -1.203  1.0   0.5   1.0   FIG. 6    3      -.5988  -1.203  1.0   0.5   1.0   FIG. 7    ______________________________________

The DC-bias values are relative to the source electrode 10. The sourceelectrode 10 is DC-biased at 0 V. In Case 1, the donor distribution hasa uniform concentration of 10¹⁶ cm⁻³. In Case 2, the donor distributionhas a local region of increased concentration--as shown in FIG. 6. InCase 3, the donor distribution has a local region of increasedconcentration--as shown in FIG. 7.

                  TABLE II(b)    ______________________________________           X.sub.SG                   X.sub.GD                           |X|.sub.max-depth                                  Y.sub.max-depth                                         Depletion Region    CASE   microns microns microns                                  microns                                         Type    ______________________________________    1      0.3     0.1     0.1    0.3    FIG. 9    2      0.3     0.1     0.1    0.3    FIG. 10    3      0.3     0.1     0.1    0.3    FIG. 11    ______________________________________

In Case 1, the shape of the depletion region 18 is of the shape shown inthe DC electron concentration contour plot of FIG. 9. In FIG. 9, thesolid-line electron concentration contour 48 has a value of 5×10¹⁵ cm⁻³,the value assigned to the depletion region edge as earlier noted. InCase 2, the shape of the depletion region 18 is of the shape shown inthe DC electron concentration contour plot of FIG. 10. In FIG. 10, thesolid line electron concentration contour 50 has a value of 5×10¹⁵ cm⁻³,the value assigned to the depletion region edge as earlier noted. InCase 3, the shape of the depletion region 18 is of the shape shown inthe DC electron concentration contour plot of FIG. 11. In FIG. 11, asolid-line electron concentration contour 52 has a value of 5×10¹⁵ cm⁻³,the value assigned to the depletion region edge as earlier noted.

The values for X_(sg), X_(gd), IXI_(max-depth), and Y_(max-depth) arerounded to the nearest 0.1 micron.

                  TABLE II(c)    ______________________________________                 a      R.sub.g                             R.sub.d                                  U      MAG    MAG          f.sub.MAX                 mi-    ohm/ ohm/ (Z.sub.0.U)                                         (Z.sub.0.MAG)                                                (Z.sub.0.MAG)    CASE  GHz    crons  cm   cm   (10.sup.-2 cm)                                         (10.sup.-2 cm)                                                (10.sup.-2 cm)    ______________________________________    1     10     0.43   600  600  9.67   <1.0   <1.0                                  (3.2)    2     <1.0   0.43   600  600  7.43   <1.0   <1.0                                  (3.25)    3     <1.0   0.43   600  600  4.24   <1.0   <1.0                                  (3.25)    ______________________________________

In Table II(c), the U, MAG, and MSG columns indicate the maximumapplicable values of these gain parameters that were observed over arange of coupling length values of 5×10⁻⁴ cm to 5×10⁻² cm, for a signalfrequency of 100 GHz. All MAG and MSG values were less than unity forthis entire range of coupling lengths. The coupling lengths for themaximum U-function values and the magnitude of the U-function valueshould be interpreted as approximate values for these parameters and thef_(MAX) frequency is approximated to within 5 GHz.

In FIGS. 61-72, the variation with frequency for the AC admittancematrix elements for the FETs in this series of "reverse" configurationTWFETs is shown. FIGS. 61-64 show the AC admittance matrix elements forCase 1, FIGS. 65-68 show the AC admittance matrix elements for Case 2,and FIGS. 69-72 show the AC admittance matrix elements for Case 3. Thecharge matrix element values have been normalized to the value of thedielectric constant of free space, to yield a dimensionless quantity. InFIGS. 73-84, the variation with frequency for the AC charge matrixelements for this series of "reverse" configuration TWFETs is shown.FIGS. 73-76 show the AC charge matrix elements for Case 1, FIGS. 77-80show the AC charge matrix elements for Case 2, and FIGS. 81-84 show theAC charge matrix elements for Case 3. In FIGS. 61-84, the notation usedfor the subscripts dg, ds, gd, and gs of the charge and admittancematrix elements is explained in the detailed description of the chargematrix calculation. The curve markers indicate the frequency values 1GHz, 40 GHz, 70 GHz and 120 GHz in sequence for each matrix elementcurve.

In Table I, Cases 5 and 6 and Table II, Cases 2 and 3, the TWFETs havelocalized regions of increased donor density. This non-uniform donordistribution provides an important advantage for these cases relative tothe other cases--the region of increased donor density will reduce themovement of the depletion region edge 20 when large values ofinter-electrode AC voltages are present in signals propagating throughthe TWFET structure. Thus, these cases can provide linear signalamplification for a signal with a larger AC voltage component than thosecases without localized regions of increased donor density. Theseadvantages of increased donor density can be extended to structures witha uniform conductivity, by increasing the donor density whilesimultaneously reducing the electron mobility so that the conductivityvalue of 1.6 Siemens/cm is maintained. The increased donor densityrestricts the movement of the depletion region edge, while theadjustment of the conductivity value allows the high frequencycharacteristics of the TWFET to remain unchanged.

It will be obvious to those of ordinary skill in the art that the ACcharge and admittance matrices of FIGS. 13-84, in combination with theseries resistance values of Table I and Table II, determine theperformance of the TWFET. A different analysis, using transverseelectric and magnetic fields, would provide the same results when thesame admittance and inductive coupling matrices are obtained andcombined with these series resistance values.

While this invention has been explained with reference to the structuredisclosed herein, it is not confined to the details set forth and thisapplication is intended to cover any modifications and changes as maycome within the scope of the following claims:

What is claimed is:
 1. A traveling wave field-effect transistor operatedat frequencies in the microwave range or above the microwave range, andhaving signals propagating therethrough generally from and to electrodesattached thereto defining a traveling wave signal direction, saidtraveling wave field-effect transistor defining a semiconductorstructure, said semiconductor structure, taken in cross section in atransverse direction at right angles to said traveling wave signaldirection, corresponding to a cross section field effect transistor,comprising:at least one gate electrode constructed in said structure, atleast one source electrode region constructed in said structure, atleast one drain electrode region constructed in said structure, adepletion region generally beneath said gate electrode wherein, in aplane transverse to said direction of signal propagation, said depletionregion defining an edge located between the gate electrode and a drainelectrode region, and means for separating the depletion region edgefrom the drain electrode region such that at some frequency saidtraveling wave field effect transistor provides, compared to said crosssection field-effect transistor, higher values for at least one of theparameters selected from the group consisting of Mason's U function,maximum available gain, and maximum stable gain.
 2. The traveling wavefield-effect transistor of claims 1, wherein a conductivity of asemiconductor of the traveling-wave field effect transistor issubstantially 1.6 Siemens/cm.
 3. The traveling wave field-effecttransistor of claims 1, wherein a conductivity of a semiconductor of thetraveling-wave field effect transistor is increased in a local areaadjacent to said drain electrode region between the depletion regionedge and the drain electrode region.
 4. The traveling wave field-effecttransistor of claims 1, further comprising:means for forming at leastone gate electrode, at least one drain electrode region, and at leastone source electrode region, means for positioning said edge of saiddepletion region wherein said edge has a first end portion locatedbetween said gate electrode and said drain electrode region and saidedge has a second end portion located distal from said gate electrode ina direction towards said source electrode region, and wherein said gateelectrode is about one micron in length.
 5. The traveling wavefield-effect transistor of claims 1, further comprising:means forforming a reverse bias between the at least one gate electrode and theat least one drain electrode region which is of a magnitude less thanthat of the reverse bias between the at least one gate electrode and theat least one source electrode region.
 6. The traveling wave field-effecttransistor of claim 1, wherein a signal is input to a drain-sourceelectrode pair and the signal is taken from a gate-source electrodepair.
 7. The traveling wave field-effect transistor of claim 1, furthercomprising at least one additional traveling wave field-effecttransistor wherein said traveling wave field-effect transistors areconfigured to operate in parallel.
 8. A traveling wave field-effecttransistor operated at frequencies in the microwave range or above themicrowave range, and having traveling wave signals propagating in adirection therethrough generally from and to electrodes attachedthereto, comprising:semiconductor structure defining a traveling wavesignal propagation direction and a transverse direction configured atright angles to said traveling wave signal propagation direction, saidstructure in cross section taken in said transverse directionperpendicular to said traveling wave signal propagation direction, saidcross section corresponding to a cross section field-effect transistor,a coupling length of said structure in said traveling wave signalpropagation direction having electrodes configured for attachingtransmission lines for an input signal and for an output signal, saidinput and output attachments at opposite ends of said coupling length,at least one gate electrode extending along said coupling length in thetraveling wave signal propagation direction, at least one sourceelectrode extending along said coupling length in the traveling wavesignal propagation direction, at least one drain electrode extendingalong said coupling length in the traveling wave signal propagationdirection, wherein a traveling wave field-effect transistor is formed,input transmission line attached to the electrodes at one end of saidcoupling length for an input signal, output transmission line attachedto the electrodes at said opposite end of said coupling length for anoutput signal, a depletion region generally beneath said at least onegate electrode , said depletion region, when viewed in a cross sectionof said semiconductor structure taken in said transverse direction,having an edge, means for positioning said edge between said at leastone gate electrode and said at least one drain electrode region, andmeans for separating the depletion region edge from the at least onedrain electrode region such that said traveling wave field effecttransistor provides, compared to said cross section field-effecttransistor, higher values for at least one of the parameters selectedfrom the group consisting of Mason's U function, maximum available gain,and maximum stable gain.
 9. A traveling wave field-effect transistoroperated at frequencies in the microwave range or above the microwaverange, and having traveling wave signals propagating in a directiontherethrough generally from and to electrodes attached thereto,comprising:semiconductor structure defining a traveling wave signalpropagation direction and a transverse direction configured at rightangles to said traveling wave signal propagation direction, a couplinglength of said structure in said traveling wave signal propagationdirection having electrodes attaching transmission lines for an inputsignal and for an output signal, said input and output attachments atopposite ends of said coupling length, at least one gate electrodeextending along said coupling length in the traveling wave signalpropagation direction, at least one source electrode extending alongsaid coupling length in the traveling wave signal propagation direction,at least one drain electrode extending along said coupling length in thetraveling wave signal propagation direction, wherein a traveling wavefield-effect transistor is formed, input transmission line attached tothe electrodes at one end of said coupling length for an input signal,output transmission line attached to the electrodes at said opposite endof said coupling length for an output signal, a depletion regiongenerally beneath said at least one gate electrode, said depletionregion, when viewed in a cross section of said semiconductor structuretaken in said transverse direction, having an edge, means forpositioning said edge between said at least one gate electrode and saidat least one drain electrode region, and means for separating saiddepletion region edge from the drain electrode region such thatnon-reciprocal inductive coupling is created along said coupling length.10. A semiconductor structure forming a traveling wave field-effecttransistor operated at frequencies in the microwave range or above themicrowave range, said structure defining at least one of gate electrodeand source and drain electrode regions, said structure having travelingwave signals propagating therethrough defining a traveling wave signaldirection and a transverse direction configured at right angles to saidtraveling wave signal direction, said electrodes and electrode regionsdefining at least two transmission lines, comprising:a depletion regiongenerally beneath said gate electrode wherein, in a plane in saidtransverse direction and at right angles to said traveling wave signaldirection, said depletion region defines an edge, and means forseparating said depletion region edge from the drain electrode regionsuch that non-reciprocal inductive coupling is created between said atleast two transmission lines of said traveling wave field effecttransistor.
 11. A traveling wave field-effect transistor operated atfrequencies in the microwave range or above the microwave range, andhaving traveling wave signals propagating in a direction therethroughgenerally from and to electrodes attached thereto,comprising:semiconductor structure defining a traveling wave signalpropagation direction and a transverse direction configured at rightangles to said traveling wave signal propagation direction, a couplinglength of said structure in said traveling wave signal propagationdirection having electrodes configured for attaching transmission linesfor an input signal and for an output signal, said input and outputattachments at opposite ends of said coupling length, at least one gateelectrode extending along said coupling length in the traveling wavesignal propagation direction, at least one source electrode extendingalong said coupling length in the traveling wave signal propagationdirection, at least one drain electrode extending along said couplinglength in the traveling wave signal propagation direction, wherein atraveling wave field-effect transistor is formed, input transmissionline attached to the electrodes at one end of said coupling length foran input signal, output transmission line attached to the electrodes atsaid opposite end of said coupling length for an output signal, adepletion region generally beneath said at least one gate electrode,said depletion region, when viewed in a cross section of saidsemiconductor structure taken in said transverse direction, having anedge, means for positioning said edge between said at least one gateelectrode and said at least one drain electrode region, means fordefining a first end portion of said edge as that end portion locatedbetween the gate electrode and a drain electrode region, and means fordefining a second end portion of said edge as that end portion distalfrom the gate electrode in a direction towards a source electroderegion, means for separating said depletion region edge from the atleast one drain electrode region, and means for adjusting said depletionregion edge so that the distance between the first end portion and thegate electrode is less than the distance between the second end portionand the gate electrode.
 12. The traveling wave field-effect transistorof claims 11 wherein a conductivity of a semiconductor of thetraveling-wave field effect transistor is substantially 1.6 Siemens/cm.13. The traveling wave field-effect transistor of claim 11 wherein saidgate electrode is about one micron in length.
 14. A semiconductorstructure forming a traveling wave field-effect transistor operated atfrequencies in the microwave range or above the microwave range, saidstructure defining at least one of gate electrode and source and drainelectrode regions, said structure having traveling wave signalspropagating therethrough defining a traveling wave signal direction anda transverse direction configured at right angles to said traveling wavesignal direction, comprising:a depletion region generally beneath saidgate electrode wherein, in a plane in said transverse direction and atright angles to said traveling wave signal direction, said depletionregion defines an edge, and means for positioning said edge between saidat least one gate electrode and said at least one drain electroderegion, means for defining a first end portion of said edge as that endportion located between the gate electrode and a drain electrode region,and means for defining a second end portion of said edge as that endportion distal from the gate electrode in a direction towards a sourceelectrode region, means for separating said depletion region edge fromthe at least one drain electrode region, and means for adjusting saiddepletion region edge so that the distance between the first end portionand the gate electrode is less than the distance between the second endportion and the gate electrode.